Photonic-assisted microwave phase shifter using a DMZM and an optical bandpass filter

Wei Li; Wen Hui Sun; Wen Ting Wang; Li Xian Wang; Jian Guo Liu; Ning Hua Zhu

doi:10.1364/OE.22.005522

1. Introduction

Microwave phase shifter has many applications such as phased-array antennas and microwave signal processing for electronic warfare systems. The conventional microwave phase shifters suffer from some limitations in terms of working bandwidth, tuning speed, and the phase shifting range. In recent years, microwave photonic phase shifter has been a hot topic due to its numerous advantages such as wide bandwidth, low loss, rapid tunability, and immunity to electromagnetic interference [1–11]. Up to now, photonic-based microwave phase shifters have been reported based on slow light in semiconductor amplifier (SOA) [3], stimulated Brillouin scattering (SBS) [4,5], vector sum [6], wavelength conversion in a semiconductor laser [7], or 2-D array of liquid crystal on silicon pixels [8]. The method based on SBS usually requires a spool of long optical fiber to stimulate the SBS effect, which makes the system bulky. For an SOA-based microwave phase shifter, several stages of SOAs have to be exploited to obtain a large phase shift. Recently, microwave photonic phase shifter has also be proposed based on bidirectional use of a polarization modulator (PolM) and a polarization-maintaining fiber Bragg grating (PM-FBG) [9] or the joint use of a PolM and a polarizer [10]. The phase shift can be tuned by adjusting the polarization direction of the polarizer [10]. However, an additional high-speed PolM may be required to achieve rapid phase tuning [11].

In this paper, we report a novel wideband 360° microwave photonic phase shifter based on a conventional dual-drive Mach-Zehnder modulator (DMZM) and an optical bandpass filter (OBPF). The fundamental microwave signal to be phase shifted is applied to one arm of the DMZM, while its frequency doubled component is fed to the other arm. The optical carrier and the sidebands on either side of the optical carrier are removed by the OBPF followed after the DMZM. In this way, only two sidebands corresponding to the fundamental microwave signal and its frequency doubled component, respectively, are left. A full 360° phase shift between the two sidebands as well as that of the fundamental microwave signal recovered in the photodetector (PD) can be achieved by simply adjusting the bias voltage of the DMZM.

2. Principle

The schematic configuration of the proposed DMZM-based microwave photonic phase shifter is shown in Fig. 1(a), which consists of a laser diode (LD), a DMZM, a microwave frequency doubler (FD), an OBPF, an erbium-doped fiber amplifier (EDFA), and a PD. Each arm of the DMZM is a phase modulator. A fundamental microwave signal and its frequency doubled component are applied to the two arms of the DMZM, respectively. The normalized electrical field on each arm of the DMZM can be expressed as

E_{a r m 1} (t) = \exp j (ω_{0} t + β_{1} \sin ω_{m} t)

E_{a r m 2} (t) = \exp j (ω_{0} t + β_{2} \sin 2 ω_{m} t + θ)

where ω₀ is the angular frequency of the optical carrier. β₁ = πV_m/V_π and β₂ = πV_2m/V_π are the phase modulation indices on the two arms of the DMZM, respectively. V_π is the half-wave voltage of the DMZM. V_m and ω_m is the amplitude and the angular frequency of the fundamental microwave signal, respectively. θ = πV_b/V_π is the phase difference between the two arms of the DMZM that can be tuned by adjusting the bias of the DMZM (V_b). Moreover, V_2m is the amplitude of the frequency doubled microwave signal. Applying Jacobi-Anger expansion to Eq. (1), we have

E_{a r m 1} (t) = \sum_{n = - \infty}^{\infty} J_{n} (β_{1}) \exp [j (ω_{0} + n ω_{m}) t]

E_{a r m 2} (t) = \sum_{n = - \infty}^{\infty} J_{n} (β_{2}) \exp [j (ω_{0} + 2 n ω_{m}) t]

where J_n(·) is the Bessel function of the first kind of order n. Under small-signal condition, only the first-order sidebands are considered as shown in Fig. 1(b). The electrical field at the output of the DMZM can be expressed as

\begin{matrix} E_{D M Z M} (t) = J_{- 1} (β_{1}) \exp [j (ω_{0} - ω_{m}) t] + J_{- 1} (β_{2}) \exp [j (ω_{0} - 2 ω_{m}) t] \exp (j θ) \\ + [J_{0} (β_{1}) + J_{0} (β_{2}) \exp (j θ)] \exp (j ω_{0} t) \\ + J_{1} (β_{1}) \exp [j (ω_{0} + ω_{m}) t] + J_{1} (β_{2}) \exp [j (ω_{0} + 2 ω_{m}) t] \exp (j θ) . \end{matrix}

As can be seen from Eq. (3), the power of the optical carrier is affected by the bias voltage of the modulator, which means that the static operation point of the modulator could be changed from linear to nonlinear region by setting the bias voltage. However, it is noted that the powers of the sidebands are independent of the bias of the modulator. Then, an OBPF is used to remove the optical carrier and the sidebands on either side of the optical carrier (e. g. the sidebands on the lower frequency side). Thus, the electrical field at the output of the OBPF is given by

\begin{matrix} E_{o u t} (t) = J_{1} (β_{1}) \exp [j (ω_{0} + ω_{m}) t] \\ + J_{1} (β_{2}) \exp [j (ω_{0} + 2 ω_{m}) t] \exp (j θ) . \end{matrix}

Equation (4) shows that the power of the resulted optical signal at the output of the OBPF is independent of the bias of the modulator. If the output signal from the OBPF is launched to the PD, the output microwave current can be expressed as

\begin{matrix} i_{m} (t) \propto E_{o u t} (t) \cdot E_{o u t}^{*} (t) \\ \propto 2 J_{1} (β_{1}) J_{1} (β_{2}) \cos (ω_{m} t + θ) . \end{matrix}

It can be seen from Eq. (5) that the fundamental microwave signal is recovered in the PD and its phase is continuously tunable by adjusting the bias voltage of the DMZM. Moreover, the amplitude of the recovered microwave signal is kept unchanged during the phase tuning.

Fig. 1 (a) Schematic configuration of the proposed DMZM-based microwave photonic phase shifter. (b) Schematic optical spectra at different locations of the system.

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3. Experiment and result

An experiment was carried out to verify the proposed scheme based on the setup shown in Fig. 1(a). The LD emitting at a wavelength of 1550.087 nm was fiber-coupled to the DMZM. The DMZM has a 3-dB bandwidth of 40 GHz and a half-wave voltage of 5.2 V. The fundamental microwave signal from a vector network analyzer (VNA) was split into two parts by a 3-dB power divider which has a bandwidth of 26.5 GHz. One part of the fundamental microwave signal was applied to the upper arm of the DMZM. The other part was frequency doubled by a microwave FD (Marki ADA-0512). The frequency doubled microwave signal was fed to the lower arm of the DMZM. The FD has an input bandwidth from 5 to 12 GHz and an output bandwidth from 10 to 24 GHz.

First of all, the microwave signal from the VNA was set at a fixed frequency of 9 GHz. The black solid line in Fig. 2(a) shows the optical spectrum when the fundamental microwave signal at 9 GHz was applied to the upper arm of DMZM while the lower arm was shorted. The bias voltage of the DMZM was 0 V. As can be seen, the second order sideband is more than 20 dB lower than the first order one. The orange dashed line in Fig. 2(a) shows the optical spectrum when the frequency doubled component at 18 GHz was applied to the lower arm of the DMZM while the upper arm was shorted in turn. Due to the limited fundamental tone suppression, the microwave signal at 9 GHz was also modulated onto the optical signal, which can be seen in Fig. 2(a). However, the power of the sideband corresponding to the fundamental tone at 9 GHz is 26 dB lower than that of the frequency doubled tone at 18 GHz.

Fig. 2 (a) The optical spectra when the fundamental microwave signal at 9 GHz and its frequency doubled component at 18 GHz was applied to the two arms of the DMZM, respectively. (b) The optical spectra before and after filtering as well as the filter response.

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The optical spectrum at the output of the DMZM when both the fundamental and frequency doubled signals were applied to the DMZM is shown in Fig. 2(b) by the blue line. Then, the OBPF was used to remove the optical carrier and the sidebands on the lower wavelength side of the optical carrier. The optical spectrum after filtering is shown in Fig. 2(b) by pink line. The optical carrier is suppressed by 52 dB and is 27.8 dB lower than the sideband corresponding to the fundamental microwave signal at 9 GHz. Since the dominant optical carrier is significantly suppressed, an optical loss of ~20 dB is introduced to the system. Thus, an EDFA was added to compensate the loss. The frequency response of the OBPF is also shown in Fig. 2(b) by green line. The PD has a 3-dB bandwidth of 15 GHz was used for square-law detection. The phase shift of the fundamental microwave signal at 9 GHz versus the bias voltage of the DMZM is shown in Fig. 3. As can be seen, the phase shift linearly increases with the bias of the DMZM over a full 360° phase tuning range when the bias voltage changes from 0 to 10.4 V (2V_π).

Fig. 3 The phase shift of the fundamental microwave signal at 9 GHz versus the bias voltage of the DMZM.

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To demonstrate the wideband phase tuning property of the proposed microwave phase shifter, the frequency of the microwave signal from the VNA was swept from 8 to 11 GHz. The phase responses of the phase shifter at different bias voltages are shown in Fig. 4(a). The VNA was calibrated at the bias voltage of 0 V. Flat phase responses were obtained over a 360° phase tuning range when the bias voltage varied from 0 to 10.4 V (corresponding to the lines from bottom to the top shown in Fig. 4(a)). The magnitude responses of phase shifter at different bias voltages corresponding to that in Fig. 4(a) are also shown in Fig. 4(b). The variation of the magnitude responses is less than 3 dB except for some ripples at the frequencies around 9.4 GHz and near 11 GHz. Actually, some ripples are also observed from the phase responses at the same frequencies as shown in Fig. 4(a). This can be attributed to the limited fundamental tone suppression of the FD used in our experiment.

Fig. 4 Measured (a) phase and (b) magnitude responses of the phase shifter when the bias voltage varies from 0 to 10.4 V (corresponding to the lines from bottom to the top shown in (a)).

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The power of the fundamental and the frequency doubled tones at the output of the FD versus the frequency of the input microwave signal is shown in Fig. 5. It can be seen that the power of the frequency doubled tone varies within 3 dB from 6 to 12 GHz. However, the suppression of the fundamental tone is worse at some frequencies. The datasheet of the FD indicates a 20 dBc typical fundamental tone suppression. However, the performance of the device was degraded due to some unknown reasons. The working bandwidth of the microwave phase shifter was chosen from 8 to 11 GHz where the suppression of fundamental tone is relatively high. However, it can be seen that the fundamental tone is still high at the frequencies around 9.4 GHz and 11 GHz. This undesired fundamental tone might be strong enough to affect the desired one if it is recovered in the PD. In this case, Eq. (4) should be rewritten as

\begin{matrix} E_{o u t 1} (t) = J_{1} (β_{1}) \exp [j (ω_{0} + ω_{m}) t] \\ + η J_{1} (β_{2}) \exp [j (ω_{0} + ω_{m}) t] \exp (j θ) \\ + J_{1} (β_{2}) \exp [j (ω_{0} + 2 ω_{m}) t] \exp (j θ) \end{matrix}

where η is the amplitude ratio between the undesired fundamental tone and the frequency doubled tone at the output of the FD. The microwave current at the output of the PD is rewritten as

i_{m 1} (t) \propto \sqrt{A^{2} + B^{2} + 2 A B \cos θ} \cdot \cos [ω_{m} t + φ]

where A = ηJ₁²(β₂), B = J₁(β₁)J₁(β₂), and φ = arctan(sinθ/(cosθ + A/B)). As can be seen from Eq. (7), the amplitude and phase of the recovered fundamental microwave signal varies with η. This explains the ripples observed in the phase and magnitude responses of the microwave phase shifter as shown in Fig. 4. If a FD with excellent fundamental tone suppression is used, a microwave photonic phase shifter with flat magnitude and phase responses could be achieved.

Fig. 5 The powers of the fundamental and the frequency doubled tones at the output of the FD versus the frequency of the input microwave signal.

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We also evaluated the stability of phase shifter by running the system in the laboratory environment and sweeping the time over 120 s. Figure 6 shows the stability of the phase shifter over a full 360° phase tuning range at a fixed frequency of 8.5 GHz when the bias voltage varies from 0 to 10.4 V. The measured results show an excellent short-term stability of the proposed microwave photonic phase shifter.

Fig. 6 The phase shift of the microwave signal at the frequency of 8.5 GHz for different bias voltage of the DMZM when time is swept over 120s.

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4. Conclusion

We have theoretically and experimentally demonstrated a novel wideband 360° microwave photonic phase shifter based on a conventional DMZM and an OBPF. The two arms of the DMZM are driven by the fundamental microwave signal to be phase shifted and its frequency doubled component, respectively. An OBPF followed after the DMZM is used to achieve carrier suppressed single-sideband modulation. A full 360° microwave phase shifter has been achieved by simply adjusting the bias of the DMZM. In our experiment, the bandwidth of the proposed microwave photonic phase shifter is limited by the bandwidth of the FD. In principle, the bandwidth of the phase shifter can be improved by using a FD with wider bandwidth. It is worth noting that the bandwidth of the DMZM should be twice that of the proposed phase shifter. Moreover, the lower boundary of the phase shifter might be limited by the roll-off performance of the OBPF.

Acknowledgments

This work was supported by the National Natural Science Foundation of China under 61377069, 61335005, 61177080, 61321063, and 61090391.

References and links

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Photonic-assisted microwave phase shifter using a DMZM and an optical bandpass filter

Abstract

1. Introduction

2. Principle

3. Experiment and result

4. Conclusion

Acknowledgments

References and links

Cited By

Figures (6)

Equations (9)

Optics Express